Electronic ballast for fluorescent lamps

ABSTRACT

A high-frequency filtered dimmable electronic ballast circuit for powering low-pressure fluorescent lamps includes an input protection device; a radio-frequency filter connected to an output of the input protection device; a rectifier connected to an output of the radio-frequency filter; a power factor correcting switching power supply device connected to an output of the rectifier; a current controlled push-pull converter connected to an output of the switching power supply device; a series-parallel resonant filter composed of a capacitor in series with two inductors having a common connection in parallel with an additional capacitor connected to an output of the push-pull converter to maintain voltage wave forms essentially sinusoidal and of constant amplitude regardless of loading; a fluorescent lamp connected to an output of the resonant filter; and an external on/off switching circuit connected directly or by an optical or magnetic insulation device to an input of the push-pull converter for renotely turning on and off the fluorescent lamp.

FIELD OF THE INVENTION

The present invention relates to the supply of power for fluorescentlamps. More specifically, the invention pertains to a high-frequencyfiltered and dimmable electronic ballast circuit for use withlow-pressure fluorescent lamps. The circuit contains an input protectionmeans, a radio-frequency filter means connected to the output of saidprotection means, a rectifier means connected to the output of saidradio-frequency filter means, a power factor correcting switching powersupply means connected to the output of said rectifier means, a currentcontrolled push-pull converter means connected to the output of saidswitching power supply means, one or more series-parallel resonantfilter means connected to the output of the said push-pull convertermeans and fluorescent lamp means connected to the output of saidresonant filter means.

BACKGROUND OF THE INVENTION AND PRIOR ART

Low-pressure fluorescent tubes, so technical solutions for theirelectronical starting and powering are well known and have been widelyused for several decades. Most of prior art solutions are designed andbuild as an ON/OFF (two-state) switch for the sake of simplicity andlow-cost: the fluorescent lamp is either switched on or switched off andtheir light input intensity cannot be dimmed neither within ranges norcontinuous.

Low-pressure fluorescent lamps and ballasts for their electricalstarting and powering are well known technology and various types offluorescent lamp and ballast combinations have been widely used forseveral decades. Most older magnetic ballasts contain passive componentsonly, provide voltage outputs in the 50 Hz to 60 Hz frequency range aresubject to start-up and running flicker and operate extremelyinefficiently. Newer electronic ballasts utilize active electronicswitching circuits, provide high-frequency voltage outputs in the 20 kHzto 50 kHz frequency range and operate more efficiently. However, mostelectronic ballasts are very simple in design, have insufficient inputprotection and radio-frequency filtering, operate at poor input powerfactors, inject large amounts of low-frequency harmonics back into theac supply, have square-wave voltage outputs, utilize feed-back controlto maintain the output voltage constant and are suitable only to operatein the ON/OFF mode (i.e.: non-dimming). Such ballasts eliminate start-upand running flicker but do not operate very reliably and, in addition,may cause severe reductions in lamp lifetimes since it is well knownthat one of the major causes of reduced lamp lifetime is the result ofsubjecting the lamp filaments to high-voltage square-wave excitation.Some of the latest types of electronic ballast designs have attemptedkeep the excitation quasi-sinusoidal by resonating the outputsquare-waves with series resonant circuits. The load impedance of thelamp in such circuits is generally an integral part of the seriesresonant circuit and different ballasts are required to powerfluorescent loads with different impedances. Output voltage is sensedand feedback is used to modulate the square-wave to maintain outputvoltage relatively constant during load impedance changes. If, duringnormal ON operation, a lamp load is removed from such a circuit, theoutput voltage across the remaining lamp and across the feedback circuitmay become very high and result in the damage of the remaining lamp, theconverter's electronic circuits and also pose a potential risk to theuser.

As an example of latest technology, U.S. Pat. No. 4,933,605, discloses ahigh frequency dimmable electronic ballast circuit for low-pressurefluorescent lamps. The circuit has a dc power supply, a converterconnected to the power supply for preparing a square-wave ac voltage, aseries-resonant output circuit with the fluorescent lamp load connecteddirectly in series (or via a transformer) with the resonant circuit.Accordingly, the lamp is an integral part of the resonance circuit usedto force the waveform of the voltage between the lamp filaments to bequasi-sine wave in shape. The ballast uses feedback control to try tokeep output voltage essentially constant and it is claimed that theballast is capable of driving more than one fluorescent lamp. Althoughthis ballast appears to provide significant advantages over most priortypes, it is designed to operate correctly only with designated types oflamps and various different ballast must be used to supply differentlamp types. In addition, since the lamps are active parts of theresonance circuit, if lamp impedances increase the peak values of thevoltage between the filaments will also tend to increase. Impedanceincreases due to aging may tend to accelerate the lamp aging process andthe failure of one lamp may cause the other to also fail. In addition,removal of a lamp during normal ON operation may result in thegeneration of very high voltages that could cause damage to theremaining lamp and/or the converter circuit and may also be dangerous tothe user.

It is the object of the present invention to provide a high-frequencydimmable electronic ballast that will operate consistently regardless ofsource and load changes and is suitable to power various types offluorescent lamps. Further, during the design process, it was recognizedthat by ensuring a constant amplitude sinusoidal output regardless ofloading and at the same time fully isolating the source from the load,high-voltage caused failures of the control circuits would be virtuallyeliminated. Accordingly, it is also the object of the present inventionto provide a high-frequency dimmable electronic ballast that does notrequire feed-back from the output in order to maintain the outputvoltage constant. Finally, it is also the object of the presentinvention to provide a high-frequency dimmable electronic ballastcapable of powering low-pressure fluorescent lamp loads while meetingall "state-of-the art" input and output requirements by providing inputand output short circuit and overtemperature protection, reducingconducted and radiated radio-frequency interference (RFI), improvingefficiency and input power factor, reducing input total harmonicdistortion (THD) and limiting no-load output peak voltages. By meetingthe above objectives this ballast will be easier and safer to use andwill operate more reliably than other types presently available.

SUMMARY OF THE INVENTION

The basic invention provides a high-frequency filtered dimmableelectronic ballast circuit for powering low-pressure fluorescent lamps,containing an input protection means, a radio-frequency filter meansconnected to the output of said protection means, a rectifier meansconnected to the output of said radio-frequency filter means, a powerfactor correcting switching power supply means connected to the outputof said rectifier means, a current-controlled push-pull converter meansconnected to the output of said switching power supply means, aseries-parallel resonant filter made up of a capacitor in series withtwo inductors with their common connection in parallel with anothercapacitor to make up a passive output filter to provide load independentsinusoidal output voltage without feedback means connected to the outputof the said push-pull converter means and a fluorescent lamp meansconnected to the output of said resonant filter means.

In a first embodiment the current-controlled push-pull convertertransformer TR2 is equipped with more than one secondary windings andmore than one series-parallel resonant filters are used to power morethan one fluorescent lamp simultaneously,

In a second embodiment an external low voltage signal is connecteddirectly to an input of the current-controlled push-pull converter inorder to make dimming possible.

In a third embodiment an external low voltage signal is connected viaoptical insulation means to an input of the current-controlled push-pullconverter in order to make dimming possible.

In a fourth embodiment an external low voltage signal is connected viamagnetic insulation means to an input of the current-controlledpush-pull converter in order to make dimming possible.

In a fifth embodiment an external ON/OFF switching circuit is connecteddirectly or via optical or magnetic insulation means to an input of thecurrent-controlled push-pull converter in order to turn the fluorescentlamps ON/OFF remotely.

Some of the novel features of the ballast design are as follows:

The ballast maintains its output voltages sinusoidal and constantregardless of input voltage and frequency variations or output loadchanges. Special series-parallel resonant output filters are used tomaintain the voltage between the filaments constant and sinusoidalwithout the need for feedback voltage control. Accordingly, the ballastsmay be used to power various types and combinations of lamps in areaswhere large input voltage and frequency variations are common.

The ballast can be used to control the light output intensity ofwhatever combination of lamps are connected to its output.

The ballast provides effective soft-starting to extend filamentlifetimes and also allows the lamps to be dimmed to very low lightoutput intensities. Isolated low-voltage windings are provided for thefilaments of each lamp to ensure that some current will flow thoroughthem and they become preheated before the voltage between them becomeshigh enough to start current flow between them. The same windings alsoensure that the peak values of the filament voltages are heldessentially constant during dimming thereby allowing the lamps to bedimmed below 10% light output intensity.

The ballast is provided with an input power factor correction circuitthat compensates for input voltage variations, minimizes input currentharmonic distortion and maximizes input power factor and ballastefficiency. This feature ensures that, even if a large number of theseballasts are connected in parallel across the same supply, the supplyand supply circuit breakers will not be adversely effected.

Input RFI filtering is provided in order to minimize conducted andradiated RFI. This feature is especially important when electronicballasts are used close to noise sensitive electronic equipment such asRF receivers and computers.

The push-pull dc-to-ac converter circuit is operated in the currentcontrol mode. This helps to prevent switching transformer core fluxunbalance and saturation which can result in increased current draw bythe switching devices and excessive ballast temperature rise.

The temperature of the switching devices is sensed and the ballast issafely shut-down if the devices become overheated. The circuit can bereset by turning the input power off then back on. This feature allowsthe ballast to be safely used at ambient temperatures in excess of 45°C.

The ballast is protected against input and output short circuit, inputvoltage transients, input undervoltage and repetitive on/off/onswitching. These features allow the ballast to operate safely especiallyin areas where supply failures and supply generated over andundervoltages are common.

The power factor correcting boost converter allows the ballast tooperate from either ac of dc input voltage. This feature allows theballasts to be used in locations where dc back-up is available andsource transfer is provided.

BRIEF DESCRIPTION OF THE DRAWINGS

Reference may now be made for the following detailed description of apreferred embodiment of an electronic ballast circuit and itsembodiments which were build up pursuant to the invention, taken inconjunction with the accompanying drawings, in which

FIG. 1 illustrates a block diagram of a basic embodiment of thehigh-frequency dimmable electronic ballast circuit according to theinvention;

FIG. 2 illustrates a more reduced form of the block diagram in FIG. 1;

FIG. 3 illustrates the first embodiment of the invention in a similarformat as FIG. 2;

FIG. 4 illustrates the second embodiment of the invention in a similarformat as FIG. 2;

FIG. 5 illustrates the third embodiment of the invention in a similarformat as FIG. 2;

FIG. 6 illustrates the fourth embodiment of the invention in a similarformat as FIG. 2;

FIG. 7 illustrates the fifth embodiment of the invention in a similarformat as FIG. 2; and

FIG. 8 illustrates the detailed circuit diagram of the electroniccircuit according to the invention.

DETAILED DESCRIPTION OF A PREFERRED EMBODIMENT

Referring now in more detail to the drawings, they illustrate a highfrequency filtered and dimmable ballast designed to supply power to oneor more low-pressure fluorescent lamps of various types and powerratings. The ballast maintains the light output of the lamps essentiallyconstant over a large range of input voltage and frequency without theuse of push-pull converter feedback control. This is accomplishedthrough the use of a power factor correction/switching regulator circuitand by resonating the isolated square-wave outputs of the push-pullconverter. The power factor correction/switching regulator circuit holdsthe input to the push-pull converter constant while the series-parallelresonant filters ensure that load changes do not effect the sinusoidalfilter outputs and thereby the lamp intensities. The ballast is designedto power the lamps in the dimming or non-dimming mode. In thenon-dimming mode, the ballast maintains lamp intensities at 100% ofrated (FIGS. 1, 2 and 3). With the addition of suitable dimming devices,connected directly or via optical or magnetic insulation to thepush-pull converter control input, the ballasts output can be varied toproduce between less-than 10% to 100% of rated lamp intensity (FIGS. 4,5 and 6). With the addition of a suitable remote switching circuit, theballast can be switched ON/OFF from a remote location by light, sound ortouch (FIG. 7) The power factor correction/switching regulator circuitalso keeps the input power factor at greater than 98% and reduces theline total harmonic distortion below 5% while ensuring overallconversion efficiency of better than 90%.

The subsystem blocks of the basic electronic ballast and theirdesignated numbers within the block diagrams in FIGS. 1 through 7 are as11 listed below.

input protection circuit (1);

RFI filter circuit (2);

full-wave rectifier circuit (3);

power factor correction/switching power supply circuit (4);

push-pull converter circuit (5);

series-parallel resonant filter circuit (6);

electronic lamp load circuit (7).

The detailed operational description of circuit operation to followrefers to the circuits as shown on FIG. 8.

In the input protection circuit, the combination of inrush currentlimiting thermistor RT2, metal oxide varistor MOV1 and input fuse F1 areintended to protect the ballast against excessive inrush current whichmay damage rectifiers D4-D7, against high-voltage transients which maydamage control components and against sustained internal short circuitswhich could result in destructive damage.

The RFI filter is made up of devices C7, C8, C9, C10, L1 and L2. Thefilter circuit is intended to reduce conducted RFI, in the range of 0.45MHz to 30 MHz generated by high-speed switching of inductive componentsinternal to the ballast, from appearing on the ac supply cabling. Sincethe internal switching circuitry is shielded by the grounded metalenclosure, reducing conducted RFI on the supply leads also results inthe virtual elimination of radiated RFI.

In the full-wave rectification circuit diodes D4 through D7 full-waverectify the 85 to 265 V_(ac) input that appears across C10. The peakvalue of the unfiltered direct voltage that appears across the input tothe power factor correction circuit is in the range of 120 to 373V_(dc).

The power factor correction circuit is made up of high-frequency seriesinductor L3, field effect transistor (FET) Q2, control circuit U2,output filter capacitor C30 and various other devices required to start,power, control and protect the power factor correction circuit. Thecircuit boosts the input direct voltage to 400 V_(dc) and keeps itconstant over the rated input voltage and frequency range. Operating Q2and the rest of the ballast circuits at 400 V_(dc) reduces current flowand minimizes power losses thus improving overall ballast efficiency. Q2is switched on at the zero crossings of the current waveform through L3.Zero current switching minimizes the reverse recovery losses inrectifiers D4 through D7 thus maximizing their conversion efficiency.The control keeps the switching frequency of Q2 relatively constantduring each input half-cycle. It varies Q2 on-time as required duringeach half-cycle to maintain the power factor correction output voltageconstant during rated line voltage and frequency changes.

The power factor correction circuit maintains the input power factorclose to unity and the input current waveform sinusoidal as follows:

    V.sub.L3 =L3(di/dt)

and since, for boost converter operation, V_(L3) =V_(inst) (t) andsince, the change in current through L3 is the same as the peak value ofthe input current (since zero current switching is used) di=I_(pk) (t),and since, L3=C1 and dt=C2 (both can be considered constant during eachhalf-cycle):

    V.sub.inst (t)=(C.sub.1 /C.sub.2)* I.sub.pk (t).

Accordingly, the current through the inductor at any time during thehalf-cycle will be in phase with the voltage across the inductor and,since the input voltage across the inductor is sinusoidal, input currentwill also be sinusoidal. Thus the circuit forces the input current totrack the input voltage thereby maintaining close to unity input powerfactor. It also ensures that the input current remains sinusoidalthereby minimizing input current harmonic distortion.

When ac voltage is applied to the input of the ballast, C27 starts tocharge through R19. When the direct voltage across C27 becomes greaterthan 16 V_(dc) (at approximately 85 V_(ac) input), the circuit starts tooperate. If the ac input voltage drops below approximately 55 V, thevoltage across C27 will drop to less than 10 V_(dc) and the undervoltagelockout circuit, internal to the control circuit U2, will disable Q2firing. Inductor L3 is provided with a secondary winding which suppliespower to U2 after startup. The high frequency output from the secondaryof L3 is rectified with D11 and the resulting direct voltage is filteredby capacitors C3 and C27 to provide ripple free control voltage Vcc toinput #7 of U2.

Output voltage is controlled by varying Q2's on-time as follows. Thecircuit's output voltage appears across the voltage divider network madeup of R15, R16 and R26. If the ac input voltage rises, the output directvoltage of the circuit will also tend to rise. This causes the voltagedrop across R26 to increase and the voltage at input #1 (VFB) of U2 toincrease. V_(FB) is amplified by a factor R24/R25 and the output of thevoltage error amplifier E/A is compared with the sawtooth waveformgenerated at input #4 (RAMP input). The on-time of Q2 is reduced tocompensate for the ac input voltage increase If the ac input voltagedrops, the ramp on-time will be increased. Accordingly Q2 will remain onfor the longest period when the ac input voltage is lowest and willremain on for the shortest period when the ac input voltage is atmaximum. The ramp reference waveform's minimum frequency is set toapproximately 30 kHz by R22 and C25. Maximum Q2 on-time is thereforelimited to approximately 33 microseconds.

U2 is provided with a slow-circuit, made up of Q17, D24, D20 and C39.This circuit minimizes output voltage overshoots and is required toprevent down-stream component damage especially if repetitivestop-starts are expected. Capacitor C39 charges through Q16'sbase-emitter junction during start-up thereby initially clamping theoutput voltage of the error amplifier low and than allowing it to risegradually. This causes the on-time of Q2 to increase slowly and thecircuit's output voltage to ramp up. If power is lost (or ac inputvoltage drops below 55 V_(ac)), diode D20 discharges C39 preparing thecircuit for the next slow-start.

Output overcurrent protection is provided by sensing the current flowthrough series resistors R40-R44. The voltage developed across theparallel combination of these resistors appears at input #2 (I_(SNS)) ofU2 and is internally connected to an overcurrent comparator. If the loadcurrent rises above the maximum expected, the comparator will disable Q2firing and, at the same time, cause a rapid rise in the charging currentto C25 thereby forcing Q2 on-time to decrease to zero. This due to thefact that charging time of C25 is set by current through R22 from input#3 (I_(SET)) and, if the voltage across R22 rises, current thorough Q13and Q14 drops while current thorough Q16 increases causing C25 to chargequicker.

The push-pull converter circuit is made up of transformer T1, switchingFETs Q3 and Q4, current mode controller U1 and the various auxiliarycircuits and components required to start, power, control and protectthe circuit. The push-pull converter circuit operates by monitoring andlimiting the current flow thorough Q3 and Q4 on a pulse-by-pulse basis.Although pulse width modulation (PWM) control is used, instead ofcomparing the error voltage to a voltage ramp reference, the pulse widthmodulation compares the error voltage (V_(e)) to an analog voltagerepresentation of the current thorough the primaries of T1 (V_(s)) Thistype of peak current sensing allows safer operation of Q3 and Q4 whileensuring that the flux flowing thorough the core of T1 is balanced.Balancing the flux flow reduces the possibility of excessive T1 heatingwhich would result in increased current flow thorough Q3 and Q4,additional heat losses and a general reduction in ballast reliability.Adjustable dead-time, between when one FET turnsoff and the otherturns-on, is provided. This ensures that Q3 and Q4 will not beconducting at the same time. In addition, double pulse suppression isused to eliminate consecutive pulsing of the FET gates. Q3 and Q4 areswitched at approximately 40 kHz and their on-time can be variedremotely to control the duration that the primary windings of T1 areenergized. The circuit incorporates power-on-reset (to block firinguntil the reference regulator's output voltage is stabilized),slowstart, current limiting and switching device temperature/outputshortcircuit protection.

Starting power is supplied to input#13 (V_(c)) of control circuit U1 viaa circuit made up of R18-1, R18-2, R17, R33, Q1, DZ1 and D8. Thestarting voltage to U1 is approximately equal to the voltage across DZ1.Running power is supplied by a set of T1 center-tapped windings (pins#1,2 and 3). The ac output of T1 is rectified by diodes D3 and D13. Theresultant direct voltage is filtered with C26. As the push-pullconverter circuit starts to operate, the voltage across C26 rises aboveDZ1 voltage and Q1 turns off. R27 is used the limit control power supplycurrent while DZ2 limits the control voltage. The running power supplyreduces transformer switching losses since it converts inductiveswitching voltage pulses into useful control power.

During startup, silicon controlled rectifier SCR1 (internal to U1) isturned on. With SCR1 on, the output of the voltage error amplifierremains low ensuring that Q3 and Q4 remain turned off. When the input dcvoltage to the circuit rises high enough, the U.V lockout circuit turnson Q8 allowing SCR1 to turn off. When SCR1 turns off, the voltage acrossC38 and the output of the voltage error amplifier rises slow-starting Q3and Q4. If during normal operation SCR1 is turned on (by the input dcvoltage dropping or as a result of protection circuit operation), C38 isdischarged and Q3 and Q4 are forced off. Slow-start is reinitiated oncethe ballast input is turned off then back on.

During non-dimming mode of operation of the output voltage controlcircuit, the ac output voltage T1 supplies to the series-parallelresonant filters and the filament windings is determined by the input dcvoltage at pin #5 of T1 and the on-time of Q3 and Q4. FET on times aredetermined by the circuit switching frequency and the output voltagesfrom the voltage and current error amplifiers in U1. The non-invertinginput of the voltage error amplifier is connected to R34 while itsinverting input looks at the voltage across R2. Both the R2 and R34voltages are constant during normal non-dimming mode operation. Thevoltage error amplifiers proportional gain is set by RI and integralgain is set by R11 and C16. The inverting input of the current erroramplifier is connected to common while its noninverting input connect tothe positive voltage side of current sensing resistors R8-1,4 via thefilter network made up of R3, L8 and C22. The current error amplifierhas a constant gain of times 3. The output of voltage error amplifier(V_(s)) depends on the voltage across R2 while the output of the currenterror amplifier (V_(s)) depends on the current flowing thoroughparalleled resistors R8-1,4. The shape of V_(s) is dependent oninductive current flowing in T1's windings while V_(e) is a steady dcvoltage level. The two signals are compared by the PWM comparator in U1and, if V_(s) rises above V_(e), the PWM comparator firing is retarded.The output of the PWM comparator sets the PWM latch. The PWM latch isused to control the remaining components of the Q3 and Q4 firingcircuit. The PWM comparator corrects for current overshoots on apulse-by-pulse basis ensuring that the peak currents thorough Q3 and Q4are kept equal. Q3 and Q4 firing frequency is set by C15, R9 and U1oscillator (OSC) at approximately of 40 kHz. The voltage across R2remains constant during the non-dimming mode of operation so that T1'soutput will remain constant as long as the dc voltage input to T1remains constant and T1 is not overloaded.

In the dimming mode, the control voltage at input connections #3 and #4can be varied with various types of remote circuits. If the controlvoltage is increased, current flow thorough the light emitting diodeinside U3 will increase, the light detecting transistor inside U3 willconduct more current and the voltage across R2 will decrease. This willcause voltage error amplifier output V to drop and Q3 and Q4 firing tobe advanced. Reducing the control voltage will, in turn, cause Q3 and Q4firing to advance. When starting the ballast. Q6 is turned onimmediately in order to ensure lamp starting with low control voltageinput. Q6 pulls the inverting (INV) input of voltage error amplifierlow. This causes Q3 and Q4 firing to advance fully for a short timedelay (as determined R39, R38, C36, DZ3, D16 and Q5). When the voltageacross C36 rises above the breakdown voltage of DZ3, Q5 turns on forcingQ6 to turnoff and Q3/Q4 on-time to be reduced.

In the circuit arrangement there are several other functional units,e.g. overtemperature and overcurrent shutdown and other protectivecircuits. Shutdown comparator operation is initiated if the voltage atinput #16 (SHUTDOWN) of U1 rises above a preset level. This can occur ifSCR2 conducts due to FET overtemperature or if the current thoroughparalleled sensing resistors R8-1,4 becomes excessive. If FETtemperature exceeds the preset level the resistance of thermistor RT1will drop and DZ4 will conduct sufficient current to trigger-on SCR2. Ifeither Q3 or Q4 conducts excessive current, D19 will conduct and againSCR2 will be turned-on. In either case, the voltage across R4 will riseabove the shutdown comparator reference level causing protectiveshutdown. SCR2 will continue to conduct until the ac input to theballast is reset. When the ac is turned back on, RT1 will ensure thatpush-pull converter circuit operation will not start until FETtemperatures drop to an acceptable level while the slow-start feature ofthe push-pull converter circuit will ensure that the FETs are quicklyturned-off again if the output is still shortcircuited.

Diodes D1 and D2 are provided to commutate current switching transientsaway from the source and drain of Q3 and Q4. Switching overvoltagesacross Q3 and Q4 are limited by the circuit composed of D9, D10, R14 andC23. O9 and D10 become forward biassed when the voltage across one orother winding exceeds the voltage across C23. As the voltage across thewinding drops, C23 discharges to the input voltage at pin #5 of T1 inreadiness for the next voltage transient.

In the current limit circuit a voltage divider network, made up of R13.R12 and R35, is connected between the regulated 5.1 V output of U1 andcommon. The voltage across R12 and R35 is sensed by input #1 (CurrentLimit Input) of U1.

This voltage determines the maximum output value of the voltage erroramplifier since if it in less forward biassed and allowing more of thereference current, I_(R), to flow thorough diode D25 thereby raisingV_(e). During startup, R35 is in the circuit so that the current limitsetpoint is raised to accommodate the expected inrush current. After adelay, Q5 turns-on and shunts R35 out of the divider reducing thecurrent limit to just above the normal operating current value.

The output circuits are composed of low-voltage heater windings, highvoltage running windings, and series-parallel resonant filters andcomponents required to start ionization.

The low-voltage windings (T1 terminals #7,8, 10, 11, 12, 13, 15 and 16)are used to ensure that each filament is supplied with sufficientcurrent to keep it warm. These windings are energized during the startof push-pull converter circuit operation and preheat the filaments. Theyensure correct starting, dimming and help to extend filament lifetimes.

The series-parallel resonant filters are connected between T1's highvoltage secondary windings (T1 terminals #8, 9 and 13, 14) and the lampfilaments. The filters are made up of C18, C19, L4, L6, C20, C21, L5 andL8. They ensure that the voltage waveforms between the filaments aremaintained essentially sinusoidal even during dimming mode operation.This results in the reduction of load generated harmonics, input power,lamp filament stress and a maximization of light output and operatingefficiency. The filters ensure that the load to the push-pull convertercircuit remains essentially capacitive therefore no-load to full-loadoperation is feasible with various types of lamps and the outputs can beopencircuited without damaging the push-pull transistors in thepush-pull converter circuit. The filter output impedances aresubstantially the same as their input impedances and, since their inputimpedances are very low (essentially short-circuits), short-circuitingtheir outputs will provide sufficient current flow in the primaries ofT1 to ensure that ballast protective circuits can operate correctly.

The lamps are started with the capacitors connected between the lampfilaments (C1 and C3). On startup, the ac voltages across the filteroutputs will tend to rise slowly. When these voltages are sufficientlyhigh, C1 and C3 start to resonate with L7 and L5 thereby providing theinitial high-voltages required to start conduction between thefilaments.

While there has been shown and described what is considered to be apreferred embodiment of the invention, it will, of coarse, be understoodthat various modifications and changes in form or detail could readilybe made without departing from the spirit of the invention. It is,therefore, intended that the invention be not limited to the exact formand detail herein shown and described, nor to anything less than thewhole of the invention herein disclosed as hereinafter claimed.

What is claimed is:
 1. A high-frequency filtered dimmable electronicballast circuit for powering low-pressure fluorescent lamps,comprising:(a) an input protection means; (b) a radio-frequency filtermeans connected to an output of the input protection means; (c) arectifier means connected to an output of said radio-frequency filtermeans; (d) a power factor correcting switching power supply meansconnected to an output of said rectifier means; (e) a current controlledpush-pull converter means connected to an output of the switching powersupply means; (f) a series-parallel resonant filter composed of acapacitor in series with two inductors having a common connection inparallel with an additional capacitor means connected to an output ofthe push-pull converter means to maintain voltage wave forms essentiallysinusoidal and of constant amplitude regardless of loading; (g) afluorescent lamp connected to an output of the resonant filter; and (h)an external on/off switching circuit means connected directly or by oneof an optical and magnetic insulation means to an input of saidpush-pull converter means for remotely turning on and off saidfluorescent lamp.
 2. An electronic ballast circuit as claimed in claim 1wherein the current-controlled push-pull converter means is atransformer equipped with more than one secondary windings means and thesecondary winding means are each connected to series-parallel resonantfilter means and these filter means are each used to power fluorescentlamp means to allow to ballast to simultaneously power more than onefluorescent lamp.
 3. An electronic ballast circuit as claimed in claim 1wherein an external low voltage signal is connected directly to an inputof the current-controlled push-pull converter means in order to makedimming of one or more fluorescent lamps possible.
 4. An electronicballast circuit as claimed in claim 1 wherein an external low voltagesignal is connected via optical insulation means to an input of thecurrent-controlled push-pull converter means in order to make dimming ofone or more fluorescent lamps possible.
 5. An electronic ballast circuitas claimed in claim 1 wherein an external low voltage signal isconnected via magnetic insulation means to an input of thecurrent-controlled push-pull converter means in order to make dimming ofone or more fluorescent lamps possible.
 6. An electronic ballast circuitas claimed in claim 1 wherein input and output overvoltage and shortcircuit protection, overtemperature protection, radio-frequencyinterference (RFI) filtering, input power factor and harmonic distortioncorrection and soft-starting circuit are provided.